Methods and apparatus for a single inductor multiple output (SIMO) DC-DC converter circuit

ABSTRACT

In some embodiments, an apparatus includes a single-inductor multiple-output (SIMO) direct current (DC-DC) converter circuit, with the SIMO DC-DC converter circuit having a set of output nodes. The apparatus also includes a panoptic dynamic voltage scaling (PDVS) circuit operatively coupled to the SIMO DC-DC converter circuit, where the PDVS circuit has a set of operational blocks with each operational block from the set of operational blocks drawing power from one supply voltage rail from a set of supply voltage rails. Additionally, each output node from the set of output nodes is uniquely associated with a supply voltage rail from the set of supply voltage rails.

CROSS-REFERENCE TO RELATED PATENT APPLICATIONS

This application is a continuation of U.S. application Ser. No.14/213,215, filed Mar. 14, 2014 entitled “Methods and Apparatus forSingle Inductor Multiple Output (SIMO) DC-DC Converter Circuit,” whichclaims priority to and the benefit of U.S. Provisional Application No.61/783,121, filed Mar. 14, 2013 entitled “Multiple Output RegulatorCircuit,” which applications are each incorporated herein by referencein their entireties.

BACKGROUND

Some embodiments described herein relate generally to systems andmethods for minimizing power consumption in integrated circuits (ICs) inembedded systems.

Embedded systems can be used in a variety of applications including, forexample, providing monitoring, sensing, control, or security functions.Such embedded systems are generally tailored to specific applications,according to relatively severe constraints on size, power consumption,or environmental survivability.

In particular, one class of embedded system can include sensor nodes,such as sensor nodes for sensing or monitoring one or more physiologicparameters. Sensor nodes are implemented as ICs and can providesignificant benefit to health care providers, such as enablingcontinuous monitoring, actuation, and logging of physiologicinformation, facilitating automated or remote follow-up, or providingone or more alerts in the presence of deteriorating physiologic status.The physiologic information obtained using such a sensor node can betransferred to other systems that can be used to help diagnose, prevent,and respond to various illnesses such as diabetes, asthma, cardiacconditions, or other illnesses or conditions.

A sensor node can provide particular value to a subject or care giver ifthe sensor node includes certain features such as, for example,long-term monitoring capability and/or wearability. A long lifetime fora sensor node without maintenance, replacement, or manual rechargingbecomes ever more important as health care costs escalate or as morecare providers attempt to transition to remote patient follow-up andtelemedicine. It is believed that generally-available sensor nodes areprecluded from widespread adoption because of a lack of extendedoperational capability or wearability.

Minimizing or reducing power consumption by employing power managementtechniques is desirable in integrated circuit (IC) design. Knowntechniques for minimizing or reducing power consumption such as, forexample, dynamic voltage scaling (DVS), where the power supply of an ICis modulated according to its performance needs, has several drawbacksin practical implementation such as the output capacitor (C_(L)) of theDC-DC converter is typically large that leads to large settling time.Additionally, the energy stored in a capacitor is typically also highand thus changing the output voltage involves energy overheads.Typically such overheads limits the rate at which the V_(DD) can bescaled and hence the amount of energy that can be saved.

Other known methods such as, for example, panoptic dynamic voltagescaling (PDVS) include drawbacks such as the use of at least three DC-DCconverters, different routing devices, switches and level-convertersthat are used to implement the PDVS technology. Such a large number ofcomponents involve the use of high circuit area and high costs forimplementation.

Accordingly, a need exists for apparatus and methods for implementingenergy efficient and cost efficient methods to minimize powerconsumption by ICs used in embedded systems.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic illustration of a known dynamic voltage scaling(DVS) circuit for a multi-core system.

FIG. 2 is a schematic illustration of a known panoptic dynamic voltagescaling (PDVS) circuit for a multi-core system.

FIG. 3 is a schematic illustration of a single-inductor multiple-output(SIMO) converter circuit, according to an embodiment.

FIG. 4A is a schematic illustration showing a SIMO converter circuithaving a high-side switch and a low-side switch, according to anembodiment.

FIG. 4B is an illustration of a timing diagram that can includerespective states of the switches shown in FIG. 4A.

FIG. 5 is a schematic illustration of a SIMO converter circuit that canbe coupled to a dynamic voltage scaling (DVS) functional block,according to an embodiment.

FIG. 6 is a graphical illustration of an inductor current correspondingto the inductor shown in FIG. 5.

FIG. 7A is illustrates a simulated inductor current.

FIG. 7B illustrates the voltage of a first terminal of an inductor.

FIG. 8A is a graphical display of a comparator circuit output that canbe obtained in response to respective comparator inputs for thecomparator circuits in FIGS. 3 and 5.

FIG. 8B is a schematic illustration of a transistor configuration thatcan be provide the comparator circuit output of FIG. 8A.

FIG. 9 is an illustrative example of respective measured output nodevoltages that can be obtained from a SIMO converter circuit shown inFIG. 5.

FIG. 10 is an illustrative example of a measured efficiency of a 0.9 VDCconverter circuit, such as provided by an output of the multi-outputSIMO converter circuit as shown in FIG. 5, plotted with respect to aload current.

FIG. 11 is a schematic block diagram that shows the implementation ofthe SIMO DC-DC converter circuit to drive a PDVS system, according to anembodiment.

FIG. 12 shows a known control scheme to generate the High Side (HS)switching control for a DC-DC converter corresponding to a portion ofthe SIMO converter circuit of FIG. 11.

FIG. 13 is a schematic illustration of the circuit for an HS controlscheme, according to an embodiment.

FIGS. 14A-H shows the behavior of the (High-side) HS control circuit ofan individual DC-DC converter circuit.

FIG. 15 shows simulation results for the ripple voltage for differentvalues of decoupling capacitors at different loads.

FIGS. 16A-B shows the simulation results of the output voltage andinductor current at light load and heavy load conditions, respectively.

FIG. 17A shows an example of the variation of ripple with the comparatorquiescent current.

FIG. 17B shows an example the condition of a comparator when output loadchanges from 100 μA to 10 mA in 10 ns.

FIGS. 18A-D shows the behavior of the (Low-side) circuit of a SIMO DC-DCconverter circuit.

FIG. 19 is a circuit diagram of a SIMO controller, according to anembodiment.

FIGS. 20A-B each shows an example of the distribution of load currentfor different scenarios on the output rails.

FIG. 21A is an illustrative example of respective measured output nodevoltages, such as can be obtained from the SIMO DC-DC converter circuitas shown in FIG. 11, according to a specified output regulationpriority.

FIG. 21B is an illustrative example of respective measured output nodevoltages, such as can be obtained from the SIMO DC-DC converter circuitas shown in FIG. 11, according to a second, different output regulationpriority.

FIG. 22A is an illustrative example of the measured output nodevoltages, such as can be obtained from the SIMO DC-DC converter circuitas shown in FIG. 11, at a heavy load as compared to the example of FIG.22B.

FIG. 22B is an illustrative example of the measured output nodevoltages, such as can be obtained from the SIMO DC-DC converter circuitas shown in FIG. 11, at a light-to-moderate load as compared to theexample of FIG. 22A.

FIG. 23A is an illustrative example of the measured efficiencies of a0.9V DC converter circuit, as can be operated stand-alone, or operatedalong with other outputs in a multi-output configuration.

FIG. 23B is an illustrative example of the measured efficiencies of a0.7 VDC converter circuit, such as can be operated stand-alone, oroperated along with other outputs in a multi-output configuration.

FIG. 23C illustrates generally an illustrative example of the measuredefficiencies of a 0.4 VDC converter circuit, such as can be operatedstand-alone, or operated along with other outputs in a multi-outputconfiguration.

FIG. 23D illustrates generally an illustrative example of the measuredefficiencies of a 0.4 VDC converter circuit, such as can be obtainedfrom the SIMO DC-DC converter circuit as shown in FIG. 11, but having alower-static-current comparator and a higher-static-current comparator.

FIG. 24 is an example of a die microphotograph of an integrated circuitthat can include at least a portion of the SIMO DC-DC converter circuitas shown in FIG. 11.

FIG. 25 is a flowchart that illustrates a method for regulating anoutput voltage using one or more of the converter circuits, according toan embodiment.

SUMMARY

In some embodiments, an apparatus includes a single-inductormultiple-output (SIMO) direct current (DC-DC) converter circuit, withthe SIMO DC-DC converter circuit having a set of output nodes. Theapparatus also includes a panoptic dynamic voltage scaling (PDVS)circuit operatively coupled to the SIMO DC-DC converter circuit, wherethe PDVS circuit has a set of operational blocks with each operationalblock from the set of operational blocks drawing power from one supplyvoltage rail from a set of supply voltage rails. Additionally, eachoutput node from the set of output nodes is uniquely associated with asupply voltage rail from the set of supply voltage rails.

DETAILED DESCRIPTION

In some embodiments, an apparatus includes a single-inductormultiple-output (SIMO) direct current (DC-DC) converter circuit, withthe SIMO DC-DC converter circuit having a set of output nodes. Theapparatus also includes a panoptic dynamic voltage scaling (PDVS)circuit operatively coupled to the SIMO DC-DC converter circuit, wherethe PDVS circuit has a set of operational blocks with each operationalblock from the set of operational blocks drawing power from one supplyvoltage rail from a set of supply voltage rails. Additionally, eachoutput node from the set of output nodes is uniquely associated with asupply voltage rail from the set of supply voltage rails.

In some embodiments, an apparatus includes a single-inductormultiple-output (SIMO) DC-DC converter circuit having a set of outputnodes, a set of comparators, and a set of switches operatively coupledto the set of comparators. The set of comparators and the set ofswitches collectively define a hysteretic-based output to control a setof output nodes, where each comparator from the set of comparators isuniquely associated with an output node from the set of output nodes,and each output node from the set of output nodes is uniquely associatedwith a circuit block from a set of circuit blocks.

In some embodiments, an apparatus includes a single-inductormultiple-output (SIMO) converter circuit, where the SIMO convertercircuit includes a set of output nodes and an inductor, and a set ofcircuit blocks, where each circuit block from the set of circuit blocksis coupled to the set of output nodes. The SIMO converter circuit canoperate over a set of time periods, where the SIMO converter circuit canprioritize a single output node from the set of output nodes for eachtime period from the set of time periods such that the prioritizedsingle output node receives current from the inductor before theremaining output nodes.

In some embodiments, an apparatus includes an embedded system thatincludes a voltage converter circuit that includes a first switch thatcan couple a first input node to a first terminal of an inductor inresponse to a control signal provided to a control input of the firstswitch. The apparatus also includes a first comparator circuit, a secondcomparator circuit, a diode coupled between a second input node and thefirst switch that is coupled to the first terminal of the inductor, anda controller circuit that can selectively couple one of the output ofthe first comparator or the output of the second comparator to thecontrol input of the switch based on a specified output regulationpriority. The controller circuit can also selectively couple a secondterminal of the inductor to one of a first output node or a secondoutput node based on the specified output regulation priority.

As used in this specification, the singular forms “a,” “an” and “the”include plural referents unless the context clearly dictates otherwise.Thus, for example, the term “a comparator” is intended to mean a singlecomparator or a combination of comparators.

An embedded system, such as a sensor node, can use multiple power supplydomains or output voltages. Such a system can include a power supplycircuit configured to provide power for various functional blocksincluded as a portion of the system. The power supply circuit outputscan be adjusted or selected, and operably coupled to respectivefunctional blocks of the system based on information about a state of anenergy source.

For example, when available energy is abundant, a functional block canbe supplied by a power supply voltage adjusted or selected to provideenhanced processing performance. Similarly, when available energy islimited, the functional block can be supplied by a power supply voltageadjusted or selected to conserve energy, perhaps at a cost of decreasedprocessing performance. In one approach, a dynamic voltage scaling (DVS)technique can be used, such as to dither or select between availablepower supply voltages in real-time based on one or more of processingdemand or the state of the energy source. DVS, however, involves severaloverheads as described above.

FIG. 1 is a schematic illustration of a known dynamic voltage scaling(DVS) circuit for a multi-core system. In FIG. 1, the DVS is for asingle V_(DD), multi-core system 100 that includes a first core 130, asecond core 140 and a third core 150. The V_(DD) is scaled according tothe performance or power needs of the IC. The DVS controller 110modulates the power supply of the IC according to its performance needs.The implementation of the DVS system shown in FIG. 1 involves severaloverheads. The output capacitor (C_(L)) of a DC-DC converter 120 isusually large; therefore it has large settling time. The energy storedon the capacitor (C_(L)) is also high; therefore changing the outputvoltage involves energy overheads. Usually these overheads limits therate at which V_(DD) can be scaled and hence the amount of energy thatcan be saved. The flexibility of DVS is also limited as individual cores130-150 do not operate at their optimal voltages. Operating each core130 or 140 or 150 with its own V_(DD) can save more power. The number ofDC-DC converters 120 used for this purpose, however, increases linearlywith the number of cores 130-150. Low drop out (LDO) and switchedcapacitor converters present on-chip options for implementing DVS formulti-core systems, albeit at lower efficiency which again limits powersavings. Panoptic dynamic voltage scaling (PDVS) is another techniquethat has been used to overcome the limitations of DVS.

FIG. 2 is a schematic illustration of a known panoptic dynamic voltagescaling (PDVS) circuit for a multi-core system. The multicore system 200includes a first core (or block) 250, a second core (or block) 260, anda third core (or block) 270. Each core 250-270 inside the multi-coresystem 200 can be connected to any of the three different V_(DD) orvoltage rails through header-switches 290. The cores or blocks 250-270can be connected to a given V_(DD) rail depending on power orperformance requirements of the IC, and each V_(DD) rail is suppliedwith voltage from their respective DC-DC converters 220-240. The PDVScontroller 210 provides voltage (or current) to the appropriate DC-DCconverters 220-240 as per the performance of the multi-core system 200.This allows the cores 250-270 to switch from one voltage to another.Using three different voltage levels, a block or core 250-270 can bemade to operate at its substantially optimal voltage by using a voltagedithering technique. Using PDVS techniques, several limitations of knownDVS circuits can be overcome. In a PDVS circuit, each block can betheoretically made to operate at its substantially optimal voltage,which results in higher overall power savings. The voltage rails in aPDVS circuit are fixed and the cores (or blocks) 250-270 are connectedto these rails depending on their throughput demands. The fixed voltagelevel in a PDVS circuit eliminates or reduces the settling time andenergy overhead costs usually present in known DVS techniques because ofthe overhead cost of the DC-DC converter. As a result, a PDVS circuitcan implement a faster rate voltage scaling technique and can save morepower. The implementation of a PDVS circuit, however, involves multipleDC-DC converters 220-240, where each DC-DC converter 220-240 feeds orsupplies each V_(DD) line. The other cost involved is higher area owingto the routing, switches and level-converters (LVL) 275-279 used in thePDVS circuit. The LVL's 275-279 are operably coupled to each other viathe system bus 280. The area overhead of the switches 290, and the LVLs275-279 is less than 15% for each core (or block), which does not amountto significant cost given the energy benefit. The cost of multiple DC-DCconverters 220-240, however, used to implement the PDVS circuit canamount to significant cost when compared to a system implementing asingle V_(DD) DVS circuit as shown in FIG. 1.

In an embedded system, functional blocks can be included as a portion ofone or more semiconductor devices having a high degree of integration.For example, one or more of a memory circuit, a general-purposeprocessor circuit, or an application-specific processor circuit can beincluded on a commonly-shared integrated circuit. Such an integratedcircuit can be referred to as a “System-on-a-Chip” or SoC. It is nowaccepted that among other things, ultra-low power (ULP) techniques canbe applied to one or more circuits included in an embedded system, suchas a sensor node. For example, a SoC can include one or more analog ordigital portions configured for sub-threshold operation, such as toconserve energy. Other techniques can be used instead of sub-thresholdoperation, or in addition to sub-threshold operation, such as power orclock gating to disable or suspend operation of specified sections ofthe system, or including adjusting a duty cycle, a clock frequency(e.g., clock throttling), or a supply parameter (e.g., supply voltagethrottling) so as to reduce power consumption.

In one approach, respective power supply voltages (e.g., respectivesupply V_(DD) “rails”) can be provided by separate power supplyregulation circuits. For example, such respective power supplyregulation circuits can include linear (e.g., dissipative) or switchingtopologies to convert energy provided by an energy source (e.g., abattery or an energy harvesting circuit) to a specified regulated outputvoltage.

In contrast, it is recognized, among other things, that separate powersupply circuits can be replaced with few or even a single multi-outputpower supply regulation circuit. Such a multi-output approach can reducea power supply circuit footprint, reduce a component count (particularlydiscrete components), and can enhance efficiency. For example, asingle-inductor multi-output (SIMO) topology can provide multiplerespective regulated output voltages, such as using a single inductor.Such a SIMO topology can be used, for example, to provide respectiveregulated output voltages to a ULP SoC, such as included as a portion ofa sensor node. Such an ULP SoC can include functional blocks configuredto operate using scalable or selectable power supply voltages based onprocessing demand or in response to information about available energy.

The three output rails for a PDVS circuit can be generated through aSIMO architecture, which is a lower cost and a high efficiency solution.To further reduce the cost and system volume, the capacitors can beintegrated. The integration of capacitors can be possible, for example,when lower capacitances are used. Use of lower sized capacitances,however, typically increases the ripple on the power supply. To mitigatethis problem, lower sized on-chip capacitances can be used. The rippleon the power supply can be reduced through a hysteretic control schemedescribed herein. Furthermore, the use of SIMO also can typically resultin higher ripple and cross regulation issues because of the changes inloads on the different V_(DD) rails. This issue can be addressed bydesigning the SIMO converter to be able to configure itself based on theload information that is available in a PDVS system as described herein.

The following describes the design of a SIMO DC-DC converter withon-chip capacitors, using the features of PDVS techniques. This designcan provide a cost efficient, energy efficient way to implement blocklevel DVS. Some embodiments described herein are practicalimplementations for PDVS and implement low cost and efficient PDVS. Theuse of SIMO can reduce the cost of multiple DC-DC converter demands forsuch embodiments. Such embodiments can provide, for example, threeoutput rails at, for example, 0.9V, 0.7V, and 0.4V, respectively, andwith a peak efficiency of 86% with integrated capacitance.

FIG. 3 is a schematic illustration of a single-inductor multiple-output(SIMO) converter circuit, according to an embodiment. The SIMO convertercircuit 300 can be included as a portion of an integrated circuit, suchas shown in the illustrative example of FIG. 24. In an example, the SIMOconverter circuit 300 can include a first switch 302 that cancontrollably couple a first terminal 306 of an inductor 304 to a firstinput node VIN1. The SIMO converter circuit 300 can include a diode 310,such as coupled between the first terminal 306 of the inductor 304 and asecond input node VIN2. A second terminal 308 of the inductor 304 can becontrollably coupled to one of a first output node VOUT1, using a firstoutput switch 314A, or a second output node VOUT2, using a second outputswitch 314B.

A control input of the switch 302 can be coupled to an output of acontroller circuit 316. The SIMO converter circuit 300 can include afirst comparator circuit 312A, such as including a first input coupledto the first output node VOUT1 (or a signal proportional to VOUT1), anda second input coupled to a first output node reference voltage VREF1.The first output node reference voltage VREF1 can be proportional to orcorrespond to a nominal or specified output voltage (e.g., a set pointor target voltage for VOUT1). Similarly, the SIMO converter circuit 300can include a second comparator circuit 312B, such as including a firstinput coupled to the second output node VOUT2 (or a signal proportionalto VOUT2), and a second input coupled to the second reference voltageVREF2.

One or more of the first comparator circuit 312A or the secondcomparator circuit 312B can include a respective threshold specified atleast in part to provide a specified respective hysteresis. For example,the hysteresis can be specified at least in part to limit a ripple of anoutput voltage provided at VOUT1 or VOUT2 by the SIMO converter 300.

The controller circuit 316 can selectively couple one of the output ofthe first comparator 312A or the output of the second comparator 312B tothe control input of the switch 302, such as based on a specified outputregulation priority. Similarly, the controller circuit 316 can includeone or more respective outputs that can controllably couple one of thefirst output nodes VOUT1 or the second output node VOUT2 to the secondterminal 308 of the inductor 304 based on a specified output regulationpriority.

A voltage provided by the energy source VS can be boosted beforecoupling to one or more of the first or second input nodes VIN1 or VIN2.The SIMO converter circuit 300 can include a “buck” topology, such as todown-convert a voltage provided by the energy source VS, to providerespective regulated output voltages such as VOUT1 or VOUT2, or one ormore other output voltages. For example, during a charging phase, afirst inductor current IL1 can be established, such as using the firstswitch 302 in response to a control signal provided by the controllercircuit 316. In an illustrative example, the inductor current can belinear assuming that the voltage provided at the first input node VIN1is roughly constant. The diode 310 can be reverse-biased during such acharging phase. During a discharge phase, the first switch 302 can beopened, and the diode 310 can become forward biased as the voltage VX atfirst terminal 306 of the inductor 304 swings negative with respect tothe second input node VIN2 (e.g., a ground or reference node), and asecond inductor current IL2 can be established, through the diode 310.Although a diode 310 symbol is shown, the diode 310 can include one ormore transistors, such as connected in a diode configuration orotherwise configured to provide a diode structure (e.g., such asprovided by a field effect transistor (FET) in a cut-off mode ofoperation, or using a FET having a gate terminal shorted to a sourceterminal).

For purposes of illustration, FIG. 3 shows two comparators 112A and112B, and corresponding two output nodes VOUT1 and VOUT2, respectively,as an example only and not a limitation. In other configurations,however, the topology shown in FIG. 3 can be structured to provide morethan two comparators and their corresponding outputs. For example, asdiscussed in the examples below, the topology of FIG. 3 can be used toprovide three or more outputs.

In the case of FIG. 3, the SIMO converter circuit 300 can be coupled toan energy source VS, such as a primary or rechargeable battery or anenergy harvesting circuit. Examples of energy harvesting circuitsinclude circuits configured to receive optical energy (e.g., aphotovoltaic circuit), a thermoelectric generator (TEG), a circuitconfigured to harvest mechanical energy or vibration (e.g., apiezoelectric circuit), or a circuit configured to receiveradiatively-coupled or magnetically-coupled operating energy (e.g., aradio-frequency receiver circuit).

In the case of FIG. 3, one or more of the first comparator circuit 312A,the second comparator circuit 312B, the controller circuit 316, thefirst switch 302, the diode 310, the first output switch 314A, thesecond output switch 314B, and one or more circuits to provide voltagereferences such as VREF1 or VREF2 can be included as a portion of acommonly-shared integrated circuit. In some instances, one or more ofthe inductor 304 and/or the energy source VS can be located off-chip.

FIG. 4A is a schematic illustration showing a SIMO converter circuithaving a high-side switch and a low-side switch, according to anembodiment. In FIG. 4A, the high-side switch is represented as SH andthe low-side switch is represented as SL. Additionally, in FIG. 4A, theSIMO converter circuit 400A can include an inductor L that can becoupled to one of a first output node VOUT1 via output switch S1, or asecond output node VOUT2 via output switch S2. The inductor L can beused in a time-division-multiplexed (TDM) manner to provide regulatedoutput voltages at the first and second output node VOUT1 and VOUT2. Forexample, VOUT1 can be coupled to a first decoupling capacitor C1 (e.g.,one or more of an off-chip or an on-chip decoupling capacitor), and caninclude a first load resistance R1. Similarly, VOUT can be coupled to asecond decoupling capacitor C2, and a first load resistance R2. Loadresistances R1 and R2 can correspond to respective functional blocks, orthe outputs VOUT1 and VOUT2 can be provided to a single functional blockin a mutually-exclusive manner, such as to provide dynamic voltagescaling (DVS) as discussed in other examples herein.

FIG. 4B is an illustration of a timing diagram that can includerespective states of the switches shown in FIG. 4A. Referring to FIGS.4A and 4B, during the initial portion of phase I, the high-side switchSH can be closed and the first output switch S1 can be closed, such asto charge the inductor as seen in the IL curve. In the latter portion ofphase I, the high-side switch SH can be opened, and the low-side switchSL can be closed, such as to discharge the inductor into the load (e.g.,into capacitor C1 and load R1 in FIG. 4A) as seen by the IL curve.During phase II, the second output switch S2 is closed, and switches SHand SL are cycled in a manner similar to phase I. However, during phaseII, the resulting inductor charge gets transferred to capacitor C2 andload R2 (as shown in FIG. 4A) as seen by the IL curve. During phase II,the output voltage VOUT1 is maintained by capacitor C1, and such anoutput voltage would generally not be expected to change significantlyfor a light load. Thus, the regulator circuit topology 400A of FIG. 4Ais a suitable option for low power application because a single inductorcan be multiplexed between multiple outputs, without causing significantvoltage drops on any one of the outputs.

FIG. 5 is a schematic illustration of a SIMO controller circuit that canbe coupled to a dynamic voltage scaling (DVS) functional block,according to an embodiment. As discussed in the examples of FIG. 3 andFIGS. 4A through 4B, a SIMO topology can be used to provide multipleregulated voltage outputs, such as using a single inductor L. In theexample of FIG. 5, the high-side switch can include a p-channeltransistor (MP), such as a metal-oxide-semiconductor field-effecttransistor (MOSFET). Use of the phrase “metal-oxide-semiconductor” doesnot imply that the gate structure of such a FET must be metallic.Instead, a polycrystalline silicon gate or other conductive material canbe used for a gate electrode included as a portion of the FET structure.

In FIG. 5, the low-side switch can include an n-channel transistor (MN).A control input (e.g., a gate) of one or more of MP or MN can be coupledto an output 520 of a SIMO controller circuit 516. For example, thecontroller circuit output 520 can be shared between MP and MN, with acontrol input 528 to MN conditioned by a timer circuit 518. For example,the timer circuit 518 can include one or more programmable or fixedcontrol circuits, such as a delay circuit 522 or an on-duration controlcircuit 524. The delay circuit 522 can provide a specified delayinitiated after MP is turned off. Similarly, the on-duration of thelow-side switch MN can be established by the on-duration control circuit524, such as to provide a specified (e.g., fixed or adjustable)on-duration, such as triggered in response to turning off MP. Thecontrol input 528 can be coupled to the SIMO controller circuit 516,such as to provide information to the SIMO controller circuit 516 thatis indicative of a conduction state of the low-side switch MN.

In some instances, the SIMO controller circuit 516 can steer theinductor current (IL) to the different output nodes (VOUT1 or VOUT2),using one or more of a first switch S1 or a second switch S2. SwitchesS1 and/or S2 can include one or more of a single transistor (e.g., ap-channel transistor) or a transmission gate structure depending on thenominal or specified output voltage for the output node.

As discussed in the example of FIG. 4A, the output nodes VOUT1 or VOUT2can include their associated decoupling capacitors C1 or C2,respectively. It is now recognized that efficiency can be enhanced andspatial volume of the SIMO converter circuit 500 can be reduced, such asusing an integrated circuit having a complementarymetal-oxide-semiconductor (CMOS) architecture. For example, small CMOSscaling (e.g., using a 65 nm process node) can provide devices (e.g.,MP, MN, S1, S2) having relatively low switching loss, allowing the SIMOconverter circuit 500 switching frequency (e.g., f_(SW)) to increase ascompared to using other technologies. As f_(SW) increases, acorresponding size of the inductor L or the decoupling capacitors C1 orC2 can decrease.

Referring back to the example of FIG. 3, when the first switch 302 opens(e.g., corresponding to MP in FIG. 5), the inductor current IL can flowthrough the diode 310, To support this inductor current IL, however, thenode VX swings to a negative voltage with respect to the second inputnode VIN2 (e.g., corresponding to REF in FIG. 5), reverse biasing thediode 310, so that the current IL can eventually decay to zero (e.g.,assuming discontinuous conduction mode (DCM)). Generally, a diode 310will have cut in voltage to turn on and thus the node VX can swing to anegative voltage of about a few hundred millivolts (mV) beforeconduction through the diode 310 is established. This can present a highresistance path from the energy source (e.g., REF node) to the inductor304. As a result, conduction loss can increase. In the example of FIG.5, to reduce such loss, a transistor MN is used. For example, thetransistor MN can be turned on briefly for the period when the inductorL is carrying current (IL) and then turned off. Generally, MN should notbe biased into conduction when the inductor current IL crosses zero asit will start discharging the charge stored on the capacitor (C1 or C2)back through the inductor L, degrading the efficiency significantly.

To avoid such discharge, MN timing can be controlled by one or more ofthe SIMO controller circuit 516 or the timer circuit 518. Generally, MNshould not be biased into conduction when MP is conducting, because sucha configuration will short VIN to REF. Secondly, MN should be biasedinto conduction almost immediately after MP is biased into cutoff (e.g.,turned off), otherwise current may flow through a body diode of MN, suchas degrading efficiency and potentially even damaging MN.

As discussed above, MN should be biased into cutoff once the inductorcurrent IL crosses zero. In one approach, such control can be achievedsuch as by sensing a voltage polarity change at node VX with respect toREF or otherwise sensing the inductor current IL. However, such sensingwould generally include using a high-speed comparator circuit, and sucha comparator circuit would consume space and energy, particularly duringconditions of light loading.

In contrast, it is recognized, among other things, that a control signalcan be generated to establish conduction of MN, such as triggered tooccur after a specified duration or otherwise in response to MP beingturned off. A pulse width of such a control signal can be small enoughso that MN turns off before inductor current IL changes for the smallestexpected load current. In this manner, MN can provide a small resistancepath for an early portion of the discharge phase of the inductor L. Tofurther reduce loss, a diode structure can be included in parallel withMN. For example, a diode structure can be provided using a secondn-channel transistor MN2. The second n-channel transistor MN2 caninclude a threshold voltage that is lower than a corresponding thresholdvoltage of MN. For example, MN2 can be referred to as an LVT transistorand can be configured to provide a gate-to-source threshold voltage, VTof, for example, about 200 mV.

While at a light load condition MN can be active (e.g., biased intoconduction) to reduce low-side conduction loss, and at high load thediode structure provided by MN2 can be forward biased for a majority ofthe inductor discharge duration. For example, in a high load condition,the current through diode structure MN2 is correspondingly higher andthus the diode structure MN2 operates in a lower resistance region, thusproviding enhanced efficiency.

In FIG. 5, or in other examples, the SIMO converter circuit 500 caninclude a DVS controller circuit 530. For example, the DVS controllercircuit 530 can be coupled to respective header switches, such as headerswitches S3 or S4, so as to select or adjust an output voltage providedto respective functional blocks, such as the DVS block 532. For example,respective blocks such as the DVS block 532 can be provided with aV_(DD) voltage selected by the DVS controller circuit 530 depending onworkload, available energy, and/or one or more other parameters. In anexample, such switching can occur in switching times of, for example,Ins or less, thus allowing dynamic or real-time control of V_(DD)voltage on an-instruction-by-instruction or task-by-task basis.

In an example, the respective outputs VOUT1 or VOUT2 are selected in amutually-exclusive manner, such as using the DVS controller circuit 530.For example, a peak load can is seen by only one output at any time,such as when all blocks are connected to that output. The DVS controllercircuit 530 can provide an output 526 to the SIMO controller circuit516, such as to provide information to the SIMO controller circuit 516indicative of an output regulation priority corresponding to a voltagescaling scheme established by the DVS controller circuit 530.

For example, a SIMO converter output attached to the highest load can beassigned a highest priority by supplying such an output (e.g., VOUT1)with IL if its voltage drops below a specified threshold as indicated bya first comparator circuit 512A. Similarly, other outputs can be cateredto, such as in order of priority. For example, a dropping VOUT2 can becharged, in response to information provided by a second comparatorcircuit 512B, if not pre-empted by VOUT1 having a higher priority.

FIG. 6 is a graphical illustration of an inductor current correspondingto the inductor shown in FIG. 5. As discussed in FIG. 5, during a firstduration A, an inductor current I_(L) can increase, such as when ahigh-side switch (e.g., MP) is biased into conduction. A duration of thecharging phase MP can be established at least in part using a comparatorcircuit configured to compare an output node voltage to a referencevoltage, such as a comparator circuit including a threshold specified atleast in part to provide a specified hysteresis as shown in the exampleof FIGS. 8A and 8B. During an early portion B of a discharge phase, alow-side switch (e.g., MN) can be biased into conduction, such as for aspecified fixed duration. The specified fixed duration can beestablished by an on-time duration control circuit, such as to enhanceefficiency during light load operation. During a late portion C of adischarge phase, a forward-biased diode structure can provide alow-resistance current path for I_(L). Respective cycles, such as thecycle shown in FIG. 6 can be repeated for respective output nodes, suchas based on a specified regulation priority, as discussed in exampleselsewhere herein.

FIG. 7A is an illustrative example of a simulated inductor current andFIG. 7B illustrates the voltage of a first terminal of an inductor. Thegraphs of FIGS. 7A and 7B can be obtained during a discharge phase, suchas corresponding to the examples of FIGS. 5 and 6 during a light loadcondition. As discussed above, in an early portion 702 of the dischargephase, a node voltage VX of a first terminal of an inductor can beclamped to a reference voltage (e.g., ground or zero volts), such as fora specified duration, using a low-side switch (e.g., MN). During a lateportion 704 of the discharge phase, a diode structure can be forwardbiased (e.g., because the node voltage VX can be negative with respectto the reference voltage).

FIG. 8A is a graphical display of a comparator circuit output that canbe obtained in response to respective comparator inputs for thecomparator circuits in FIGS. 3 and 5. A delay in comparator response canbe used to establish an on-time duration, such as for a high-side switchcontrolled at least in part using the comparator circuit output (e.g., ap-channel transistor that can become conductive when the comparatoroutput is low and can become inhibited from conduction when thecomparator output is high).

A first input of the comparator can be coupled to an output node voltageor a voltage proportional to the output node voltage (e.g., VOUTMONITOR). A second input of the comparator can be coupled to a referencevoltage, VREF, such as corresponding to a target or nominal outputvoltage. The comparator circuit can include a specified hysteresis,which can be used at least in part to limit or otherwise establish aripple of an output voltage provided by the SIMO converter circuit ofFIG. 3 or 5. In an example, the hysteresis window can be defined usingan upper threshold VTH, and a lower threshold VTL, such as can bespecified relative to the reference voltage VREF.

For example, when VOUT<VTL, the high-side switch can be turned on, whichcan charge an inductor, which then can increase VOUT. As VOUT crossesVTH at T1 (e.g., VOUT>VTH), the high-side switch can be turned off, suchas allowing a decoupling capacitor or other energy storage device tosupply the load. As VOUT drops below VTL, at T2 (e.g., VOUT<VTL), thehigh-side switch can again be turned on, unless preempted by anotheroutput based on a specified regulation priority. In this manner, both aswitching frequency and an on-duration of the high-side switch can bemodulated in response to changing loads. For example, a switchingfrequency can be lower and a pulse width of the on-duration of thehigh-side switch can be shorter at lighter loads. A tradeoff can existbetween switching frequency and ripple magnitude. For example, if higherripple can be tolerated, changing the hysteresis to lower the switchingfrequency can improve efficiency but can also increase ripple. Inanother example, in an application including a ULP SoCs, more ripplescan be tolerated because such an ULP SoC can include a relatively lowoperating clock rate.

FIG. 8B is a schematic illustration of a transistor configuration thatcan be provide the comparator circuit output of FIG. 8A. In an example,transistors MN1 and MN2 can establish a differential pair andtransistors M1 through M4 can establish an active load of the comparatorcircuit. M1 and M2 can be sized similarly, and M3 and M4 can be largerin area such as to provide higher drive strength than M1 and M2, such asto provide hysteresis.

For example, when VOUT is lower than VREF, most of the current can flowthrough MN1, and thus the comparator output OUT is low. As VOUTincreases, M2 becomes more strongly biased into conduction. When VOUT isabout equal to VREF, the drive of MN2 is about equal to MN1, however M3has higher drive strength than M1 and so MN2 needs more current to pullOUTB down. This means that VOUT must exceed VREF by a margincorresponding to VTH in FIG. 8A, to drive OUTB low. Similarly, VOUT mustbe less than VREF by a margin corresponding to VTL in FIG. 8A, to driveOUT low.

FIG. 9 is an illustrative example of respective measured output nodevoltages that can be obtained from a SIMO converter circuit shown inFIG. 5. In FIG. 9, the y-axis represents voltage and the x-axisrepresents time. Such a SIMO converter circuit can be configured togenerate three respective output voltages of approximately 0.9 VDC(e.g., VOUT1), approximately 0.7 VDC (e.g., VOUT2), and approximately0.4 VDC (e.g., VOUT3), by using an input voltage of about 1V or more.

FIG. 10 is an illustrative example of a measured efficiency of a 0.9 VDCconverter circuit, such as provided by an output of the multi-outputSIMO converter circuit as shown in FIG. 5, plotted with respect to aload current. In FIG. 10, the y-axis represents efficiency as apercentage, and the x-axis represents load current (e.g., in units ofmicroamps). An efficiency at 1002 approaches approximately 86% in thiscase.

FIG. 11 is a schematic block diagram that shows the implementation ofthe SIMO DC-DC converter circuit to drive a PDVS system, according to anembodiment. First, the discussion associated with FIG. 11 relates to theoverall system architecture and the SIMO control scheme. Second, adescription of the hysteretic control scheme to reduce the size of thecapacitors is provided. Finally, the SIMO circuit with a combination ofa PDVS load is discussed with respect to cross-regulation and higherripple associated with the SIMO architecture.

The SIMO DC-DC converter circuit 1100 includes a SIMO controller 1105and a set of output nodes 1115A-C. The PDVS circuit 1130 includes a PDVScontroller 1131 and a PDVS block 1132. The PDVS circuit 1130 includesmultiple PDVS blocks. The PDVS circuit 1130 is operatively coupled tothe SIMO DC-DC converter circuit 1100, where the PDVS circuit 1130 has aset of operational blocks 1132; each operational block from the set ofoperational blocks can draw power from one supply voltage rail from aset of supply voltage rails 1117A-C. Additionally, each output node fromthe set of output nodes is uniquely associated with a supply voltagerail from the set of supply voltage rails. For example, the output node1115A is associated with the voltage rail 1117A, the output node 1115Bis associated with the voltage rail 1117B, and the output node 1115C isassociated with the voltage rail 1117C.

The SIMO DC-DC converter circuit 1100 can include a first comparator1112A and a second comparator 1112B (a third comparator 1112C is alsoshown in FIG. 11), where the first comparator 1112A can receive a firstbias current (labeled “Ref 0.9V”) and produce a control signal 1120A(the control signals are labeled as 1120A-D) to select a first outputnode from the plurality of output nodes when the first output nodeexperiences a first load (e.g., a PDVS block 1132). Similarly the secondcomparator 1112B can also receive a second bias current (labeled “Ref0.7V”) and produce a control signal 1120B to select the first outputnode from the set of output nodes when the first output node experiencesa second load lower than the first load. The second bias current is lessthan the first bias current. Additionally, a power consumption of thesecond comparator 1112B can be less than the power consumption of thefirst comparator 1112A when the SIMO DC-DC converter circuit isoperative. The efficiency of the SIMO DC-DC converter circuit 1100 ishigher when the second comparator 1112B produces the control signal toselect the first output node than when the first comparator 1112Aproduces the control the signal to select the first output node.

The second comparator 1112B can also be placed in an off mode with apower consumption less than a power consumption during an operativemode, when the first comparator 1112A produces the control signal toselect the first output node. Similarly, the first comparator 1112A canbe placed in an off mode with a power consumption less than a powerconsumption during an operative mode, when the second comparator 1112Bproduces the control signal to select the first output node. In someinstances, the SIMO DC-DC converter circuit 1100 and the PDVS circuit1130 can be included within an integrated circuit (IC), where the SIMODC-DC converter circuit 1100 can prioritize a single output node fromthe set of output nodes within a time period to indicate whichoperational blocks from the set of operational blocks (e.g., PDVS block1132) of the PDVS circuit 1130 connect to which supply voltage railsfrom the set of supply voltage rails during the time period.

The SIMO DC-DC converter circuit 1100 includes a set of switches S1-S3operatively coupled to the set of comparators 1112A-1112C, where eachcomparator from the set of comparators 1112A-1112C is uniquelyassociated with a switch from the set of switches S1-S3. Morespecifically, comparator 112A is associated with switch S1, comparator112B is associated with switch S2, and comparator 112C is associatedwith switch S3. Additionally, each comparator from the set ofcomparators 1112A-1112C is associated with a lower hysteresis thresholdfrom a set of lower hysteresis thresholds and an upper hysteresisthreshold from a set of upper hysteresis thresholds, and each comparatorfrom the set of comparators 1112A-1112C can produce a pulse having awidth based on the hysteresis thresholds such that the uniquelyassociated switch is controlled in response to that pulse.

Additionally, each comparator from the set of comparators 1112A-1112Creceives a bias current from a set of bias currents and a feedbacksignal from the output node for that comparator 1112A or 1112B or 1112C,where at least one bias current from the set of bias currents differsfrom the remaining bias currents from the set of bias currents. The setof comparators 1112A-1112C and the set of switches S1-S3 cancollectively select an output node from the set of output nodes1115A-1115C based on (1) a condition of each output node from the set ofoutput nodes 1115A-1115C, and (2) a relative priority of each outputnode from the plurality of output nodes 1115A-1115C.

The SIMO DC-DC converter circuit 1100 that includes a SIMO controller1105, (hysteretic) comparators 1112A-1112C, buck DC-DC converter 1140and provides three output nodes 1115A-1115C at supply power rails1117A-1117C. As described above, the PDVS circuit 1130 includes a PDVScontroller 1131 and a PDVS block 1132. The comparators 1112A-1112C, theSIMO controller 1105 and the DC-DC buck converter 1140 implement thecontrol loop to provide the output voltages to the output nodes1115A-1115C. The SIMO controller 1105 steers the inductor current(I_(L)) to different supply power rails 1117A-1117C through switchesS1-S3. The (hysteretic) comparators 1112A-1112C compares each supplypower rail 1117A-1117C to its reference voltage and provides a digitaloutput. The (hysteretic) comparators 1112A-1112C also control theswitching turn-on time of the selected comparator 1112A-1112C which willbe described in greater detail herein. The (hysteretic) comparators1112A-1112C control the regulation and switching of each supply powerrail 1117A-1117C. These comparators 1112A-1112C can be disabled when aparticular supply power rail 1117A-1117C is not needed in the systemshown in FIG. 11. The digital output of the comparator 1112A-1112C isreceived at the SIMO controller 1105 along with the priority signal fromthe PDVS controller 1130. These signals are used to generate controlsignals for the DC-DC buck converter 1140 as well as assign switchingsequence for the switches S1-S3. The priority signal from PDVScontroller 1131 represents or indicates the current loading scenario oneach supply power rail 1117A-1117C. The SIMO controller 1105 uses thispriority signal to set the switching priority and assigns highest andlowest priority to switches S1-S3. For example, if the 0.9V supply powerrail is heavily loaded, the priority is set for this supply power railand in the event that 0.9V comparator 1112A output goes low, the SIMOcontroller 1105 starts steering inductor current I_(L) into the 0.9Vsupply power rail. In case when all the three rails are heavily loaded,then this switching configuration can result in higher ripple or crossregulation. All the supply power rails, however, are not loadedsimultaneously in a PDVS circuit 1130. As a result, higher ripple istypically not expected on the other supply power rails. Additionally,the system in FIG. 11 allows for SoC decoupling of the variouscapacitances that can save system level volume and cost.

The design goal of the PDVS-SIMO architecture as shown in FIG. 11involves several parameters. First, the PDVS-SIMO architecture should beable to support lower sized capacitance with small ripple. Second, thePDVS-SIMO architecture should be able to provide high efficiency.Finally, the static power consumption of the PDVS-SIMO architectureshould be small. The design presented in FIG. 11 can be used to reducethe overall system power and cost. The switching time of the converter1140 is controlled by the comparator 1112A-1112C which helps in reducingthe additional control circuit, which in turn reduces static powerconsumption of the system. Two approaches can be used to reduce thedimensions of the passives in the converter 1140. First, use of 65 nmadvanced process nodes where the smaller CMOS technology can enablefaster switching frequency for the converter 1140 which can enable lowersizes for inductor and capacitors. Second, the new hysteretic controlscheme can be implemented that can further bring down the size of thecapacitors to nF range.

FIG. 12 shows a known control scheme to generate the High Side (HS)switching control for a DC-DC converter corresponding to a portion ofthe SIMO converter circuit of FIG. 11. Typically, two approaches, forexample, can be used. In the first approach, a fixed delay is generatedand the turn-on time of the HS switch (MP) is controlled by the fixeddelay. This delay generates the current in the inductor I_(L) and HSswitching is enabled, followed by Low Side (LS) switching. The HScontrol determines the amount of energy getting transferred in eachcycle. The second approach for HS control uses current sensing of theinductor L and can be more desirable for higher output power switchingconverters. This scheme, however, can have some inaccuracies and higherenergy overhead, which makes it typically unsuitable for low-energylower voltage systems. Additionally, fixing the current of the inductorL involves use of a high capacitance value decoupling capacitor toreduce the ripple. The inductor current I_(L) gets stored on thecapacitor (shown in FIG. 11) for the cases of light load. This can causehigher ripple if the capacitor is small.

FIG. 13 is a schematic illustration of the circuit for an HS controlscheme, according to an embodiment. The circuit 1300 uses a hystereticcomparator 1305 to control the HS switch. The hysteresis of thecomparator 1305 and the delay of the comparator 1305 set the ripple onthe supply power rail. The high side transistor, M_(P) is turned ON whenV_(O) goes below Th_(LO) of the hysteretic comparator, where Th_(HI) andTh_(LO) are the higher and lower thresholds of the comparator 1305,respectively, with hysteresis given by, Th_(HI)−Th_(LO). The inductor Lstarts charging the output rail and its current starts ramping up. OnceV_(O) crosses Th_(HI), M_(P) is disabled and inductor current I_(L)starts discharging through to the output capacitor.

FIGS. 14A-H shows examples of the behavior of the (High-side) HS controlcircuit of an individual DC-DC converter circuit. Specifically, FIG. 14Aand FIG. 14B are illustrative examples of a simulated inductor currentthat can be obtained for respective load currents, such as using theSIMO DC-DC converter circuit topology as shown in FIG. 11. As shown inFIG. 14A, a higher load current (e.g., 10 mA) increases both a peakinductor current I_(L) and a duration of the inductor current I_(L)pulse. Similarly, as shown in FIG. 14B, a lower load current (e.g., 1mA) correspondingly reduces a peak inductor current I_(L) and a durationof the inductor current I_(L) pulse.

FIG. 14C is an illustrative example of a simulated output node voltageVOUT of a SIMO DC-DC converter circuit topology, such as shown in theexample of FIG. 11. FIG. 14D is an illustrative example of a simulatedgate-to-source voltage VGSMP that can be provided to a high-side (HS)p-channel transistor, such as to obtain the output node voltage shown inthe illustrative example of FIG. 14C. As shown in FIG. 14D, durationswhere VGSMP is low correspond to duration where a high-side switch(e.g., MP) is conducting.

FIG. 14E is an illustrative example of a simulated output node voltageVOUT of a SIMO DC-DC converter circuit topology, such as shown in theexample of FIG. 11. FIG. 14F is an illustrative example of a simulatedgate-to-source voltage VGSMP that can be provided to a p-channeltransistor, such as to obtain the output node voltage shown in FIG. 14E.The examples of FIGS. 14C and 14D show a load condition that isrelatively heavier than the examples of FIGS. 14E and 14F. In thelighter-load examples of FIGS. 14E and 14F, an on-duration of thehigh-side switch can be relatively shorter than in the example of FIGS.14C and 14D, and a duration between on-pulses can be longer. FIG. 14 Gis similar to FIG. 8A and illustrates the hysteresis thresholds. FIG.14H is a schematic illustration of a portion of the DC-DC convertercircuit.

Said in another way, in general, FIGS. 14A-H shows that at the lightload condition the voltage at the output rises quickly, because thecurrent drawn from the output capacitor is low and most of the currentis used to charge the capacitor. Therefore, the hysteresis of thecomparator 1405 (as seen in FIG. 14H) sets a lower inductor currentI_(L) and ensuring lower ripple. If the load current increases, the risetime of the output voltage increases, as a result M_(P) is on for longerduration of time (because higher current is drawn out of the output).This increases the peak current I_(L) in the inductor L. The circuit ofFIG. 14H adapts itself with the output load. This scheme makes theripple on the supply power rail less dependent on the output capacitor(not shown in FIGS. 14A-H).

FIG. 15 shows simulation results for ripple voltage for different valuesof decoupling capacitors at different loads. The different graphsplotted represent different load currents. The ripple voltage variesfrom 30-60 mV for sample values of 0.8V V_(DD) and 1.2V V_(in). A 4.3 nFof output capacitor gave approximately 5% ripple on the rail. This valueof capacitance is significantly smaller than the typical decouplingcapacitors (μF range) used on supply power rails in a DC-DC converter.At these values, the capacitance can easily be integrated on-chip. Asignificant percentage of the capacitance can come from the parasiticcapacitance of the cores connected to these supply power rails.

The HS control scheme described herein can support loads up to 50 mA andcan operate the SIMO converter circuit in both continuous conductionmode (CCM) and discontinuous conduction mode (DCM). At light loadcondition the SIMO converter circuit goes into DCM. The HS turn-on timeis used to charge the inductor. After LS control cycle the inductorcurrent I_(L) goes to zero. However, V_(O) goes below Th_(LO) later thanLS cycle owing to the light load condition. The HS control scheme startsagain, once V_(O) goes below Th_(LO). Operating in DCM at light loadhelps in achieving acceptable efficiency as well as controlled ripple.The SIMO converter circuit operates in CCM at high load condition. Athigh load condition, V_(O) goes below Th_(LO) before inductor currentgoes to zero and hence conducts continuously. Operating in CCM helps intargeting heavy load condition for the SIMO converter circuit. FIG. 16shows the simulation results of the output voltage and inductor currentat light load and heavy load conditions, respectively. FIG. 16A showsthe simulation results of the output voltage and inductor current atload current of 0.4 mA. FIG. 16B shows the simulation results of theoutput voltage and inductor current at load current of 40 mA. Theresults of FIG. 16A-B shows that the SIMO converter circuit operates inCCM at high load conditions (FIG. 16B) and in DCM at light loadconditions (FIG. 16A).

The static power consumption of the HS circuit is dictated by the powerconsumption of the comparator (e.g., comparator 1112A-1112C in FIG. 11).More power can be saved by lowering the comparator quiescent current.The performance of the comparator, however, also controls the amount ofripple seen on the supply power rail. For example, if the comparator hashigher delay (lower power), then the response of the comparator to thechanges at the output voltage will be slower, which will result inelevated ripple. FIG. 17A shows an example of the variation of ripplewith the SIMO comparator quiescent current. The ripple decreases withincreased quiescent current. The ripple becomes constant after 25 μA ofcomparator current. After this point, the hysteresis of the comparatorand output capacitor controls the amount of ripple. In the example ofFIG. 17A, a selection of 25 μA quiescent current was made for thecomparator of 0.9V and 0.7V supply power rails and two comparators wereused for 0.4V supply power rail. One comparator had a 3 μA quiescentcurrent while other comparator had a 100 nA quiescent current. This wasdone to lower the static power consumption of the converter for the lowpower mode, where all the cores in PDVS system can be connected to 0.4V.The other two converters can be disabled while 0.4V rail will supply theV_(DD) with higher ripple.

Hysteresis of the comparator is also used in determining the peakinductor current, which is used in determining the overall efficiency ofthe SIMO converter particularly at light load conditions. At highervalue of inductor current, conduction loss increases and thus decreasingthe efficiency, while at lower value of the inductor current, theswitching loss reduces the efficiency. The inductor current iscontrolled through the hysteresis in the comparator. At high loadconditions the losses are governed by the load current as the SIMOconverter operates in continuous conduction mode. The SIMO comparatorand the output capacitor also affect for the transient behavior of theSIMO converter. FIG. 17B shows an example of the condition of a SIMOcomparator when output load changes from 100 μA to 10 mA in 10 ns. Asshown in FIG. 17B, the SIMO converter can continue to regulate even fora fast change in load condition. Owing to the small size of the outputcapacitor, however, a very sharp change in output load to high load suchas 40-50 mA can result in overshoot or undershoot at the output rail.The SIMO converter uses a few μs to recover under such conditions. Thishappens because the converter goes from DCM to CCM in very short timewhich can result in higher inductor current build up and can causehigher ripple. If the load changes slowly, however, then higher rippleis not seen on the supply power rail.

FIGS. 18A-D shows the low side (LS) control of an individual SIMO DC-DCconverter circuit, according to an embodiment. Generally, FIGS. 18A-Dshow that the low side control is implemented to keep the LS switch onfor fixed time. For the rest of the time, LS conducts as a diode. The LSswitch is implemented with a low threshold voltage (LVT) device. As aresult, the diode does not contribute to significant loss. A knownscheme for LS control implements a zero detection comparator. For lightload condition and lower inductor current, the performance for zerodetects comparator is very high. For higher performance, static currentin the comparator becomes higher which adds significant overhead on theDC power consumption of the converter. The scheme to implement fixeddelay turn on time for LS eliminates this overhead with smaller penaltyin efficiency. FIG. 18A is similar to FIG. 7A, FIG. 18B is similar toFIG. 7B, and FIG. 18C is similar to FIG. 6. FIG. 18D is a schematicillustration of a portion of the SIMO DC-DC converter circuit.

FIG. 19 is a circuit diagram of a SIMO controller, according to anembodiment. In FIG. 19, the SIMO controller 1905 is operably coupled tothree comparators 1912A-1912C. The HS control is enabled when any one ofthe three (hysteretic) comparator 1912A-1912C output goes high. At thistime LS is disabled. Once HS is disabled, LS turns on for a given pulsewidth (as shown in FIG. 18). Switches S1-S3 are the SIMO switches for0.9V, 0.7V, and 0.4V rails redrawn here from FIG. 11. The priorityselection selects between c1 and c2 and correspondingly S1 and S2. Forexample, if 0.7V rail has higher priority, then c2 gets connected to p1and c1 to p2. Similarly, b1 gets connected S2 and b2 gets connected S1.Only one switch out of S1, S2, and S3 is turned on at any given time.Priority selection plays a role in selecting a particular switch. Ifmore than one supply power rail is below its Th_(LO), then the switchcorresponding to the supply power rail with higher priority is turnedon. A higher priority is assigned to the rail with the higher load. Thisdesign suits PDVS systems well. If one supply power rail is heavilyloaded, then other supply power rails are going to be lightly loaded inPDVS. Additionally, because load information is well known in the PDVScircuit (through the header switch connection), priority can be assignedcorrectly.

In this scheme, the supply power rail with higher load gets servicedfirst which prevents the supply power rail from having higher ripple.The supply power rail with lower load discharges slowly and can beserviced in the meantime. The SIMO controller 1905 can undergo highercross regulation in the case when difference in load becomes too largeamong supply power rails. This happens in CCM mode of operation. If onesupply power rail is heavily loaded with, for example, 40-50 mA current,while other rail is lightly loaded at, for example, 10-100 μA current,then the lightly loaded supply power rail can charge up because of thehigher current present in the inductor. One method to overcome thislimitation is to short both the terminals of the inductor, which resultsin energy loss. Hence, the extra current is dumped on the 0.4V railwhich has the least priority. In the event when voltage goes higher, aclamp can be used to control the voltage at 0.4V supply power rail.

FIGS. 20A-B shows the distribution of load current for differentscenarios on the output supply power rails. FIG. 20A shows the caseswhen most of the cores are connected to the 0.9V rail and FIG. 20B showsthe case when most of the cores are connected to the 0.7V supply powerrail, respectively. FIGS. 20A-B provides an insight that if one V_(DD)supply power rail is heavily loaded (case when most of the cores areconnected to that V_(DD)) other V_(DD) supply power rails are lightlyloaded. This is a unique characteristic of PDVS circuits that can beused to the advantage of SIMO converter design. This feature can be usedto address the issue of cross-regulation in SIMO converters.Cross-regulation arises in SIMO converters usually operating in CCM,when the changes in load current of one of the supply power rail resultsin the change of output voltage of another supply power rail. Thishappens largely because of the load transients on supply power rails ina system. Often, SIMO converters are over-designed to address the worstcase load transients to address cross-regulation. For the well-definedloading configurations in a system implementing PDVS, the loadtransients are typically known in advance. It is also typically knownthat which, of the three supply power rails is going to be heavilyloaded and which is lightly loaded at any point in time. Thisinformation can be easily obtained by scrutinizing the number of cores(or type of cores) getting connected to given supply power rail at agiven time because the information is already known or needed toconfigure the header switches in PDVS (see FIG. 2). Such information isused in the SIMO converter design to set the priority of the switchingsupply power rails. For example, in some instances, if a SIMO isdesigned to provide 0.9V, 0.7V and 0.4V supply power rails and 0.9Vsupply power rail is heavily loaded, then 0.9V rail is set on thepriority. In such instances, the 0.9V supply power rail is regulatedfirst and followed by the 0.7V and 0.4V rails. If loading information isnot known priority cannot be set correctly and higher cross-regulationcan be seen on the supply power rails. In order to maintain the supplypower rails within the specified error, often higher value of capacitorsare used with capacitances in μF range. Here, however, the SIMOconverter circuit can be used in conjunction with the PDVS circuit tosignificantly lower the value of decoupling capacitor used in theproposed SIMO converter.

In PDVS circuits, the abrupt change of load due to DVS control isdeterministic. If most of the cores switch to 0.7V rail from the 0.9Vrail, the PDVS controller will indicate this via a signal to SIMOconverter, which will prioritize the 0.7V rail. The feed-forwardinformation can be used to dynamically reassign the priority thatcontrols cross regulation due to abrupt load transition. This eliminatescomplex feedback schemes that will also reduce efficiency at light loadconditions.

FIG. 21A is an illustrative example of measured output node voltages,such as can be obtained from the SIMO DC-DC converter circuit 1100 asshown in FIG. 11, according to a specified output regulation priority.Such a SIMO DC-DC converter circuit can be configured to generate threerespective output voltages of approximately 0.9 VDC (e.g., VOUT1),approximately 0.7 VDC (e.g., VOUT2) and approximately 0.4 VDC (e.g.,VOUT3), such as using an input voltage of about 1V or more.

FIG. 21B is an illustrative example of measured output node voltages,such as can be obtained from the SIMO DC-DC converter circuit 1100 asshown in FIG. 11, according to a second, different output regulationpriority. Such a SIMO DC-DC converter circuit can be configured togenerate three respective output voltages of approximately 0.9 VDC(e.g., VOUT1), approximately 0.7 VDC (e.g., VOUT2) and approximately 0.4VDC (e.g., VOUT3), such as using an input voltage of about 1V or more.

Cross regulation, which can refer to increased ripple on one supplypower rail because of an abrupt change of load on another supply powerrail, can be a major issue for multi-output regulation circuit. It isrecognized that using a DVS circuit, a change in load can bedeterministic. For example, if respective blocks switch from using the0.9 VDC supply power rail to using the 0.7 VDC supply power rail, suchas in response to the DVS controller circuit, a power supply regulatorcontroller circuit can correspondingly adjust an output regulationpriority to prioritize the 0.7 VDC output, or even to ignore or disablethe 0.9 VDC output. In the illustrative example of FIG. 21A, the 0.9 VDCoutput is at a relatively lower regulation priority than the 0.7 VDCrail. Because of the difference in output regulation priority, at 2102,the 0.9 VDC rail can drop significantly, such as providing a ripple thatis 30 mV larger in magnitude than a corresponding output as shown inFIG. 21B, where the 0.9 VDC rail is assigned to limit ripple to lessthan about 40 mV. Hence, proper assignment of priority to the differentoutput rails based on load can help reduce ripple effects. In the FIGS.21A and 21B, the vertical scale is about 100 mV per division.

FIG. 22A is an illustrative example of the measured output nodevoltages, such as can be obtained from the SIMO DC-DC converter circuit1100 as shown in FIG. 11, at a heavy load as compared to the example ofFIG. 22B. Such a converter circuit can be configured to generate threerespective output voltages of approximately 0.9 VDC (e.g., VOUT1),approximately 0.7 VDC (e.g., VOUT2) and approximately 0.4 VDC (e.g.,VOUT3), such as using an input voltage of about 1V or more. The heavyload of the example of FIG. 22A includes respective output currents ofabout 10 mA on the 0.9 VDC output, about 1 mA on the 0.7 VDC output, andabout 1 mA on the 0.4 VDC output. In this example, the converter circuitefficiency is 86% with a measured ripple of about 40 mV or less.

FIG. 22B is an illustrative example of the measured output nodevoltages, such as can be obtained from the SIMO DC-DC converter circuit1100 as shown in FIG. 11, at a light-to-moderate load as compared to theexample of FIG. 22A. Such a converter circuit can be configured togenerate three respective output voltages of approximately 0.9 VDC(e.g., VOUT1), approximately 0.7 VDC (e.g., VOUT2) and approximately 0.4VDC (e.g., VOUT3), such as using an input voltage of about 1V or more.The light-to-moderate load of the example of FIG. 22B includesrespective output currents of about 10 mA on the 0.9 VDC output, about100 μA on the 0.7 VDC output, and about 100 μA on the 0.4 VDC output. Inthis example, the converter circuit efficiency is 86% and in the region2202, the converter circuit can provide sufficient inductor currentI_(L) dumps to maintain the 0.4 VDC rail at a level above 0.4 VDC (e.g.,to avoid a reset condition). FIGS. 21A-B and FIGS. 22A-B together showthat the high load efficiency is 86% and the low load efficiency is 62%.

FIG. 23A is an illustrative example of the measured efficiencies of a0.9V DC converter circuit, such as can be operated stand-alone, oroperated along with other outputs in a multi-output configuration, suchas can be obtained from the SIMO DC-DC converter circuit 1100 as shownin FIG. 11. In a stand-alone configuration, a peak efficiency of the0.9V DC output (rail) can be approximately 88%. The efficiency wasmeasured with the load current on the 0.9V DC output (or rail) in theSIMO configuration, where the load on the 0.7V and 0.4V rail was 100 μA.

FIG. 23B is an illustrative example of the measured efficiencies of a0.7 VDC converter circuit, such as can be operated stand-alone, oroperated along with other outputs in a multi-output configuration, suchas can be obtained from the SIMO DC-DC converter circuit 1100 as shownin FIG. 11. In a stand-alone configuration, a peak efficiency of the0.7V DC output (rail) can be approximately 82%. The efficiency wasmeasured with the load current on the 0.7V DC output (or rail) in theSIMO configuration, where the load on the 0.9V and 0.4V rail was 100 μA.

FIG. 23C illustrates generally an illustrative example of the measuredefficiencies of a 0.4 VDC converter circuit, such as can be operatedstand-alone, or operated along with other outputs in a multi-outputconfiguration, such as can be obtained from the SIMO DC-DC convertercircuit 1100 as shown in FIG. 11. In a stand-alone configuration, a peakefficiency of the 0.4V DC output (rail) can be approximately 61%. Theefficiency was measured with the load current on the 0.4V DC output (orrail) in the SIMO configuration, where the load on the 0.9V and 0.7Vrail was 100 μA.

FIG. 23D illustrates generally an illustrative example of the measuredefficiencies of a 0.4V DC converter circuit, such as can be obtainedfrom the SIMO DC-DC converter circuit 1100 as shown in FIG. 11, buthaving a lower-static-current comparator and a higher-static-currentcomparator. In the example of FIG. 23D, a peak efficiency of the 0.4V DCconverter circuit, using a comparator circuit consuming about 3 μA, canbe approximately 68%. A peak efficiency of a 0.4V DC converter circuit,using a comparator circuit consuming about 100 nanoamperes (nA), can beapproximately 61%. Hence, the efficiency was measured on the 0.4V DCoutput (or rail) that was operated in low and high quiescent currentmode with the 0.9V and 0.7V rails disabled.

FIG. 24 is a die microphotograph of an integrated circuit that caninclude at least a portion of the SIMO DC-DC converter circuit 1100 asshown in FIG. 11. Such a circuit can include switches and on-chipdecoupling capacitors, such as fabricated using a 65 nm CMOS processnode. Neglecting an off-chip inductor, a total area of the convertercircuit is about 1 mm×2 mm. The capacitors for the design areimplemented using NMOS capacitors. The value of the capacitance is 4.3nF for 0.9V and 0.7V rails and 2.3 nF for 0.4V rail. The capacitancescontribute to ˜1 μA of standby current because of the gate oxideleakage. The total area of the converter is 2 mm². The area of thecontrol circuits is 0.03 mm².

FIG. 25 is a flowchart that illustrates a method for regulating anoutput voltage using one or more of the converter circuits, according toan embodiment. At 2502, a voltage at a first output node can be comparedto a first output node reference, such as using a comparator circuit asshown and described in the examples of FIG. 3, 5, 8A, 8B, or 11.Similarly, at 2504, a voltage at a second output node can be compared toa second output node reference, such as using a second comparatorcircuit. At 2506, one of the output of a first or second comparator canbe coupled to a first switch, such as based on a specified outputregulation priority.

At 2508, a first input node can be coupled to a first terminal of aninductor, such as using a first switch in response to a signal providedat the control input of the first switch. At 2510, a second terminal ofthe inductor can be coupled to one of the first output node or thesecond output node, such as based on the specified output regulationpriority. In an example, one or more of the first comparator circuit orthe second comparator circuit can include a respective thresholdspecified at least in part to provide a specified respective hysteresis,such as discussed in relation to the examples of FIG. 8A or 8B andelsewhere.

It is intended that some of the methods and apparatus described hereincan be performed by software (stored in memory and executed onhardware), hardware, or a combination thereof. For example, the controlsoftware on the cell phone can be performed by such software and/orhardware. Hardware modules may include, for example, a general-purposeprocessor, a field programmable gate array (FPGA), and/or an applicationspecific integrated circuit (ASIC). Software modules (executed onhardware) can be expressed in a variety of software languages (e.g.,computer code), including C, C++, Java™, Ruby, Visual Basic™, and otherobject-oriented, procedural, or other programming language anddevelopment tools. Examples of computer code include, but are notlimited to, micro-code or micro-instructions, machine instructions, suchas produced by a compiler, code used to produce a web service, and filescontaining higher-level instructions that are executed by a computerusing an interpreter. Additional examples of computer code include, butare not limited to, control signals, encrypted code, and compressedcode.

Some embodiments described herein relate to a computer storage productwith a non-transitory computer-readable medium (also can be referred toas a non-transitory processor-readable medium) having instructions orcomputer code thereon for performing various computer-implementedoperations. The computer-readable medium (or processor-readable medium)is non-transitory in the sense that it does not include transitorypropagating signals per se (e.g., a propagating electromagnetic wavecarrying information on a transmission medium such as space or a cable).The media and computer code (also can be referred to as code) may bethose designed and constructed for the specific purpose or purposes.Examples of non-transitory computer-readable media include, but are notlimited to, magnetic storage media such as hard disks, floppy disks, andmagnetic tape; optical storage media such as Compact Disc/Digital VideoDiscs (CD/DVDs), Compact Disc-Read Only Memories (CD-ROMs), andholographic devices; magneto-optical storage media such as opticaldisks; carrier wave signal processing modules; and hardware devices thatare specially configured to store and execute program code, such asApplication-Specific Integrated Circuits (ASICs), Programmable LogicDevices (PLDs), Read-Only Memory (ROM) and Random-Access Memory (RAM)devices.

While various embodiments have been described above, it should beunderstood that they have been presented by way of example only, and notlimitation. Where methods and steps described above indicate certainevents occurring in certain order, the ordering of certain steps may bemodified. Additionally, certain steps may be performed concurrently in aparallel process when possible, as well as performed sequentially asdescribed above. Although various embodiments have been described ashaving particular features and/or combinations of components, otherembodiments are possible having any combination or sub-combination ofany features and/or components from any of the embodiments describedherein.

For example, while many of the embodiments described herein arediscussed in the context of a cell phone, other types of mobilecommunication devices having a commercial radio can be used such as, forexample, a smart phone and a tablet with wireless communicationcapabilities. Similarly, while many of the embodiments described hereinare discussed in the context of sending and receiving data packets, anytype of data unit may be applicable including data cells and dataframes, depending upon the applicable communication standard.

What is claimed is:
 1. A method, comprising: receiving informationrelated to one or more of a processing power demand and an availableenergy level of an integrated circuit (IC) that includes asingle-inductor multiple-output (SIMO) direct current-direct current(DC-DC) converter circuit and a panoptic dynamic voltage scaling (PDVS)circuit operatively coupled to the SIMO DC-DC converter circuit, theSIMO DC-DC converter circuit having a plurality of output nodes, eachoutput node from the plurality of output nodes being uniquely associatedwith a supply voltage rail from a plurality of supply voltage rails, andthe PDVS circuit having a plurality of operational blocks; and operatingthe PDVS circuit such that each operational block from the plurality ofoperational blocks draws power from one supply voltage rail from theplurality of supply voltage rails.
 2. The method of claim 1, furthercomprising: receiving, at a first comparator of the SIMO DC-DC convertercircuit, a first bias current and producing a control signal to select afirst output node from the plurality of output nodes when the firstoutput node experiences a first load; and receiving, at a secondcomparator of the SIMO DC-DC converter circuit, a second bias currentand producing a control signal to select the first output node from theplurality of output nodes when the first output node experiences asecond load lower than the first load; the second bias current beingless than the first bias current, a power consumption of the secondcomparator being less than a power consumption of the first comparatorwhen the SIMO DC-DC converter circuit is operative, an efficiency of theSIMO DC-DC converter being higher when the second comparator producesthe control signal to select the first output node than when the firstcomparator produces the control the signal to select the first outputnode, the second comparator configured to be placed in an off mode witha power consumption less than a power consumption of the secondcomparator during an operative mode, when the first comparator producesthe control signal to select the first output node, and the firstcomparator configured to be placed in an off mode with a powerconsumption less than a power consumption of the first comparator duringan operative mode, when the second comparator produces the controlsignal to select the first output node.
 3. The method of claim 1,wherein the SIMO DC-DC converter circuit and the PDVS circuit areincluded within an integrated circuit (IC), the method furthercomprising: prioritizing within a time period a single output node fromthe plurality of output nodes of the SIMO DC-DC converter circuit basedon which operational blocks from the plurality of operational blocks ofthe PDVS circuit connect to which supply voltage rails from theplurality of supply voltage rails during the time period.
 4. The methodof claim 1, further comprising: receiving, at a first comparator of theSIMO DC-DC converter circuit, a first bias current and producing acontrol signal for a first output node from the plurality of outputnodes; and receiving, at a second comparator of the SIMO DC-DC convertercircuit, a second bias current and producing a control signal for asecond output node from the plurality of output nodes; the second biascurrent being less than the first bias current, an output voltage forthe second output node being less than an output voltage for the firstoutput node, a power consumption of the second comparator being lessthan a power consumption of the first comparator when the SIMO DC-DCconverter circuit is operative.
 5. The method of claim 1, wherein: theSIMO DC-DC converter circuit includes a plurality of comparators and aplurality of switches operatively coupled to the plurality ofcomparators; each comparator from the plurality of comparators isuniquely associated with a switch from the plurality of switches; eachcomparator from the plurality of comparators is uniquely associated witha bias current from a plurality of bias currents, at least one biascurrent from the plurality of bias currents differing from the remainingbias currents from the plurality of bias currents; and each switch fromthe plurality of switches is associated with an output node from theplurality of output nodes; the method further comprising: collectivelysending, from the plurality of comparators, a control signal to controleach switch from the plurality of switches such that current is sent toeach output node from the plurality of output nodes based on thatcontrol signal; and collectively regulating, by the plurality ofcomparators and the plurality of switches, voltages on each output nodefrom the plurality of output nodes.
 6. The method of claim 1, furthercomprising: receiving, at each comparator from a plurality ofcomparators of the SIMO DC-DC converter circuit, a bias current from aplurality of bias currents and a feedback signal from an output node forthat comparator and from the plurality of output nodes, at least onebias current from the plurality of bias currents differing from theremaining bias currents from the plurality of bias currents; andcollectively selecting, by the plurality of comparators and a pluralityof switches operatively coupled to the plurality of comparators, anoutput node from the plurality of output nodes based on (1) a conditionof each output node from the plurality of output nodes, and (2) arelative priority of each output node from the plurality of outputnodes.
 7. The method of claim 1, wherein: the SIMO DC-DC convertercircuit includes a plurality of comparators and a plurality of switchesoperatively coupled to the plurality of comparators, each comparatorfrom the plurality of comparators being uniquely associated with aswitch from the plurality of switches; and each comparator from theplurality of comparators is associated with a lower hysteresis thresholdfrom a plurality of lower hysteresis thresholds and an upper hysteresisthreshold from a plurality of upper hysteresis thresholds; the methodfurther comprising: producing, at each comparator from the plurality ofcomparators, a pulse having a width based on hysteresis thresholdsassociated with that comparator such that the switch uniquely associatedwith that comparator is controlled in response to that pulse.
 8. Themethod of claim 1, further comprising: prioritizing within a time perioda single output node from the plurality of output nodes of the SIMODC-DC converter circuit based on which operational blocks from theplurality of operational blocks of the PDVS circuit connect to whichsupply voltage rails from the plurality of supply voltage rails duringthe time period; and limiting, by a plurality of comparators of the SIMODC-DC converter circuit and a plurality of switches operatively coupledto the plurality of comparators, a ripple of an output voltage within apredefined range for each output node from the plurality of outputnodes, the ripple of the output voltage for the single output node beingless than a ripple of the output voltage for each remaining output nodefrom the plurality of output nodes.
 9. A method, comprising:collectively limiting, by a plurality of comparators of asingle-inductor multiple-output (SIMO) direct current-direct current(DC-DC) converter circuit and a plurality of switches operativelycoupled to the plurality of comparators, a ripple of an output voltagewithin a predefined range for a plurality of output nodes of the SIMODC-DC converter circuit; and collectively defining, by the plurality ofcomparators and the plurality of switches, a hysteretic-based voltageoutput threshold to control the plurality of output nodes, eachcomparator from the plurality of comparators being uniquely associatedwith an output node from the plurality of output nodes, and each outputnode from the plurality of output nodes being uniquely associated withan operational block from a plurality of operational blocks of apanoptic dynamic voltage scaling (PDVS) circuit.
 10. The method of claim9, further comprising: receiving, at a first comparator of the pluralityof comparators, a first bias current and producing a control signal toselect a first output node from the plurality of output nodes when thefirst output node experiences a first load; and receiving, at a secondcomparator of the plurality of comparators, a second bias current andproducing a control signal to select the first output node from theplurality of output nodes when the first output node experiences asecond load lower than the first load; the second bias current beingless than the first bias current, a power consumption of the secondcomparator being less than a power consumption of the first comparatorwhen the SIMO DC-DC converter circuit is operative, an efficiency of theSIMO DC-DC converter circuit being higher when the second comparatorproduces the control signal to select the first output node than whenthe first comparator produces the control the signal to select the firstoutput node, the second comparator configured to be placed in an offmode with a power consumption less than a power consumption of thesecond comparator during an operative mode, when the first comparatorproduces the control signal to select the first output node, and thefirst comparator configured to be placed in an off mode with a powerconsumption less than a power consumption of the first comparator duringan operative mode, when the second comparator produces the controlsignal to select the first output node.
 11. The method of claim 9,further comprising: collectively selecting, by the plurality ofcomparators and the plurality of switches, an output node from theplurality of output nodes based on (1) a condition of each output nodefrom the plurality of output nodes, and (2) a relative priority of eachoutput node from the plurality of output nodes.
 12. The method of claim9, further comprising: receiving, at each comparator from the pluralityof comparators, a bias current from a plurality of bias currents and afeedback signal from an output node for that comparator, at least onebias current from the plurality of bias currents differing from theremaining bias currents from the plurality of bias currents; andcollectively selecting, by the plurality of comparators and theplurality of switches, an output node from the plurality of output nodesbased on (1) a condition of each output node from the plurality ofoutput nodes, and (2) a relative priority of each output node from theplurality of output nodes.
 13. The method of claim 9, wherein: eachcomparator from the plurality of comparators is associated with a lowerhysteresis threshold from a plurality of lower hysteresis thresholds andan upper hysteresis threshold from a plurality of upper hysteresisthresholds, at least one lower hysteresis threshold differs from theremaining lower hysteresis thresholds from the plurality of lowerhysteresis thresholds, and at least one upper hysteresis thresholddiffers from the remaining upper hysteresis thresholds from theplurality of upper hysteresis thresholds.
 14. The method of claim 9,wherein: the SIMO DC-DC converter circuit and a plurality of capacitorsare included within an integrated circuit (IC), each output node fromthe plurality of output nodes is uniquely associated with a capacitorfrom the plurality of capacitors, and a size of each capacitor from theplurality of capacitors is less than a size of a capacitor for a SIMODC-DC converter circuit without hysteretic-based voltage outputthresholds.
 15. A method, comprising: operating, at a single-inductormultiple-output (SIMO) direct current-direct current (DC-DC) convertercircuit, over a plurality of time periods, the SIMO DC-DC convertercircuit having a plurality of output nodes and an inductor; andprioritizing a single output node from the plurality of output nodes foreach time period from the plurality of time periods such that thatsingle output node receives current from the inductor before theremaining output nodes from the plurality of output nodes.
 16. Themethod of claim 15, further comprising: receiving, at a first comparatorof a plurality of comparators of the SIMO DC-DC converter circuit, afirst bias current and producing a control signal to select a firstoutput node from the plurality of output nodes when the first outputnode experiences a first load associated with a panoptic dynamic voltagesealing (PDVS) circuit that includes a plurality of operational blocks,each operational block from the plurality of operational blocks beingcoupled to the plurality of output nodes; and receiving, at a secondcomparator of the plurality of comparators, a second bias current andproducing a control signal to select the first output node from theplurality of output nodes when the first output node experiences asecond load associated with the PDVS circuit lower than the first load;placing the second comparator in an off mode with a power consumptionless than a power consumption of the second comparator during anoperative mode, when the first comparator produces the control signal toselect the first output node; and placing the first comparator in an offmode with a power consumption less than a power consumption of the firstcomparator during an operative mode, when the second comparator producesthe control signal to select the first output node; the second biascurrent being less than the first bias current, a power consumption ofthe second comparator being less than a power consumption of the firstcomparator when the SIMO DC-DC converter circuit is operative, and anefficiency of the SIMO DC-DC converter circuit being higher when thesecond comparator produces the control signal to select the first outputnode than when the first comparator produces the control signal toselect the first output node.
 17. The method of claim 15, wherein: thesingle output node is a first single output node, the SIMO convertercircuit and a plurality of operational blocks are included within anintegrated circuit (IC), each operational block from the plurality ofoperational blocks being coupled to the plurality of output nodes; themethod further comprising: activating a second single output node fromthe plurality of output nodes of the SIMO DC-DC converter circuit withina time period such that the operational block uniquely associated withthat second single output node is activated to regulate within the ICfor that time period.
 18. The method of claim 15, further comprising:collectively selecting, by a plurality of comparators of the SIMO DC-DCconverter circuit and a plurality of switches operatively coupled to theplurality of comparators, an output node from the plurality of outputnodes based on a current of a load operatively coupled to a plurality ofoperational blocks, each operational block from the plurality ofoperational blocks being coupled to the plurality of output nodes. 19.The method of claim 15, further comprising: limiting, by a plurality ofcomparators of the SIMO DC-DC converter circuit and a plurality ofswitches operatively coupled to the plurality of comparators, a rippleof an output voltage for the plurality of output nodes within apredefined range.